My friend Bela, who is an experienced DHT user and very fond of the 801a, asked me if I could help in developing an LTSpice model for the venerable 801a. I have several 10/10Y but unfortunately no 801a at hand to trace so I used the data sheet curves which are a bit challenging due to resolution/scaling of the characteristic curves:

801a model from data sheet curves

I had to compromise the model fit as if you want accuracy in A1 region the matching is less than perfect in A2. Still, the model is not perfect but is very good for simulation purposes.

The interesting thing was to add the grid current model for A2 operation:

A2 grid current model

Well, looking to test this model, I chose the following operating point:

Test point for the model

The data sheet grid space is a bit challenging, but looking at the model in LTSpice:

Testing the 801a in LTSpice

The following slight difference can be observed:

801a test in LTSpice.

There’s about a 10% difference. Not bad considering the normal variance you’d expect in the valve anyway.

Here is the 801a-A2-model. Let me know your feedback:

**** 801A CURVES ** Composite DHT with Advanced Grid Current **************
* Created on 01/18/2016 19:48 using paint_kit.jar 2.9
* www.dmitrynizh.com/tubeparams_image.htm
* Plate Curves image file: 801a curves.png
* Data source link: RCA data sheet
*
* Model developed by Ale Moglia (c) 2016
* valves@bartola.co.uk
*
* www.bartola.co.uk/valves
*
*———————————————————————————-
.SUBCKT DHT-801A-A2 1 2 3 4 ; P G K1 K2
+ PARAMS: CCG=4.5P CGP=6P CCP=1.5P RFIL=6
+ MU=8 KG1=2984.1 KP=124.5 KVB=2083.2 VCT=-7.04 EX=1.29
+ VGOFF=1.17 IGA=0.00099 IGB=0.03 IGC=-18.3 IGEX=1.9
* Vp_MAX=800 Ip_MAX=250 Vg_step=10 Vg_start=100 Vg_count=20
* Rp=4000 Vg_ac=55 P_max=20 Vg_qui=-48 Vp_qui=300
* X_MIN=39 Y_MIN=33 X_SIZE=786 Y_SIZE=492 FSZ_X=1418 FSZ_Y=656 XYGrid=false
* showLoadLine=n showIp=y isDHT=y isPP=n isAsymPP=n showDissipLimit=y
* showIg1=n gridLevel2=y isInputSnapped=n
* XYProjections=n harmonicPlot=n harmonics=y
*———————————————————————————-
RFIL_LEFT 3 31 {RFIL/4}
RFIL_RIGHT 4 41 {RFIL/4}
RFIL_MIDDLE1 31 34 {RFIL/4}
RFIL_MIDDLE2 34 41 {RFIL/4}
E11 32 0 VALUE={V(1,31)/KP*LOG(1+EXP(KP*(1/MU+V(2,31)/SQRT(KVB+V(1,31)*V(1,31)))))}
E12 42 0 VALUE={V(1,41)/KP*LOG(1+EXP(KP*(1/MU+V(2,41)/SQRT(KVB+V(1,41)*V(1,41)))))}
RE11 34 0 1G
G11 1 31 VALUE={(PWR(V(32),EX)+PWRS(V(32),EX))/(2*KG1)}
G12 1 41 VALUE={(PWR(V(42),EX)+PWRS(V(42),EX))/(2*KG1)}
RCP1 1 34 1G
C1 2 34 {CCG} ; CATHODE-GRID
C2 2 1 {CGP} ; GRID=PLATE
C3 1 34 {CCP} ; CATHODE-PLATE
RE2 2 0 1G
EGC1 81 0 VALUE={V(2,31)-VGOFF} ; POSITIVE GRID THRESHOLD
GG1 2 31 VALUE={0.5*(IGA+IGB/(IGC+V(1,31)))*(MU/KG1)*(PWR(V(81),IGEX)+PWRS(V(81),IGEX))}
EGC2 82 0 VALUE={V(2,41)-VGOFF} ; POSITIVE GRID THRESHOLD
GG2 2 41 VALUE={0.5*(IGA+IGB/(IGC+V(1,41)))*(MU/KG1)*(PWR(V(82),IGEX)+PWRS(V(82),IGEX))}
.ENDS
*$

I’ve been using my sound card and Pete Millett’s interface for testing. However, the limitation on high frequency response is due to the sound card. A cheap, but yet effective option, is to use the Chinese digital oscilloscope Instrustar ISDS2062B which comes with built-in DDS signal generator with sweep capability. With this little piece of technology, you can sweep up to 10Mhz. The resolution is 12bit, not great for FFT but good enough for a FR analysis. For frequencies above 20kHz, you can use this device to look into things. For FFT and THD analysis, I will keep using the sound card and audio interface as usual.

When I test the device, I found that the DDS output didn’t have the stones to drive loads at HF. Therefore I went back to my workshop and built the following sweep driver. This was based on the great SSM2019 that my friend Mogens sent me:

Sweep buffer based on SSM2019

The circuit is very simple. It’s operated from a pair of 9V batteries. So far, I’ve tested with the 20dB (actually 19dB gain due to 1kΩ resistor at hand) and 0dB gain modes.

The response is good enough for my purposes:

I can get 1.5Mhz @ 19dB or 3Mhz @ 0dB gain HF response which is great. I now need to test it again with a real load.

Introduction: DHT madness

I’m not going to dwell on DHT sound. I’d rather say that if you’re looking for a stellar DHT candidate, the 4P1L beats them all. It’s dirty cheap, reliable and sounds amazing. You can go any route you like, it’s your own decision of course. However if you’re looking for a 100% DHT amp to build, here is an interesting example for your consideration.

I’ve tried 4P1L in many topologies. The advantage of its low filament requirements is that you can implement it in filament bias and simplify the circuit significantly.

The gyrator driver using the PCB I designed recently, can be used to avoid iron and have an excellent first stage and make this Russian Amp in steroids: 4P1L driving 4P1L.

One minor caveat around 4P1L in excess. I have found (as well as many others) that if you use too many 4P1L stages (e.g. 4P1L line stage driving a 4P1L-4P1L amp) then it will sound a bit harsh in the treble. i suspect this may be explained due to the H3 component level when triode-strapped. I’d rather limit the number of 4P1L stages to two. You’ve been warned.

The other great thing about the 4P1L is that is quite consistent between samples and easy to match pairs. Also in PSE mode you can drive it to full tilt with only 20Vrms and achieve up to 5W in class A1 with a pair of valves.

The circuit

This is a variation of what a I have built in the past. I haven’t built exactly this amplifier, however I’m sure that it will perform as well as my previous versions.

The topology of this amp is dead simple. It has a first stage (driver) mu-follower with a gyrator load. The driver is coupled by a capacitor (C2) into the output stage (PSE) formed by a pair of 4P1L. These are also with filament bias. All DHTs have a Rod Coleman regulator which will allow to extract the best of the DHT sound with no compromise:

4P1L PSE 5W max amplifier

The beauty of this design is that you need only one HT (330V) and of course the filament supplies which are unavoidably the complexity of any DHT circuit. The gyrator is based on a J310 lower FET to allow the first 4P1L to run at a very healthy 30mA and ensure there are no slew rate issues in driving the output stage. The top FET (e.g. DN2540) will carry the heavy load of dropping the voltage about 100V and therefore will need a heatsink to dissipate the 3W of heat. The filament bias of the first stage will burn about 10W on the resistor so you ought to have a 20W wire wound part here. The output stage filament bias is more demanding with 16W burnt on each resistor. I used Dale’s 10Ω/5Ω/3Ω parts to make the equivalent filament bias. These Dale NOS wire wound sound great but check the DIYAudio 4P1L PSE thread as there are several recommendations around filament resistors.

How much output power? Well, with a 3K transformer, you will easily get 3.5W or so. That is plenty of power for a single ended stage if you have high efficiency speakers. The maximum achievable is 5W. However, the H3 spectrum is higher due to grid current. I prefer the sound of the 4P1L PSE up until 3 or 4W. To get the most out of the 4P1L you can drive it hard to 300V, it will work fine:

4P1L PSE Zaa=3K 5W output

If we bias the 4P1L pair at 300V / 30mA each valve, then the bias needs to be around -26V. The filament resistor is either 40Ω or 35Ω depending in which leg of the filament you place the filament bias due to the voltage dropped by the 4P1L filament.

I used an LL1623/90mA but the LL1623/60mA version should do fine (in fact, any nice 3K or 5K SE output transformer should work fine here). If you want more power you can use a 4P1L triplet and distortion will drop 1% and you can achieve higher output power. If you want less distortion and you don’t need 5W peak then you can wire the output transformer in 5K:8 and you will get 2.5W @1.5%. I  liked this impedance ratio, sounded really nice and clean.

With 18.5Vrms required to get the maximum output power and a 4P1L driver with gain of approximately 9, you need 2Vrms to get all the juice out of this amp. No further preamp is needed if you’re driving this amp with a conventional CD or similar source. A volume control is needed at the input though.

Well, hope you’ve enjoyed this post and hopefully I have temped you to explore yourself into the DHT world and build a nice 4P1L PSE amp!

Well, it was obvious I couldn’t leave last post as it was. There is an option to change the driver for a different valve. You can use a C3m (low gain in triode mode which is ideal here), a C3g, E180F/E280F. 6S45P or my loved 6E5P (or 6E6P) as the driver. Not longer a 100% DHT, but a nice option for sure. The 6E5P is extremely linear, good driver, with a nice gain (μ=30) in triode – perhaps more than enough for a 4P1L stage and would help in avoiding additional filament supplies.

The 6E5P has curves not dissimilar to the 4P1L as no further distortion cancelation can be seen. Here is the updated schematic if you’re interested in playing with:

6E5P driver for the 4P1L PSE Amplifier

Again, the gyrator PCB can be easily used to simplify the build of this amp. The 6E5P is not driven hard, but at a nice current of 20mA which makes the driver operate in a linear region (and with good sound) with just a pair of red LEDs. The nearly 30dB of gain will make this amp to be very sensitive. The 5W can be easily achieved with 1Vpp, so you will need to have an attenuator, no preamp needed clearly.  The 6E5P will drive an 300B nicely here which needs the voltage gain, not like the 4P1L.

As you can see, there are plenty of option to try on this 4P1L PSE amplifier.

From Russia with Love

East-West Divide, Copyright by Justmeans

The interesting combination to explore from our previous designs is to mix some western valves like 01a into the Russian parade.

The result would be quite interesting, as the sound of the 01a has proven to be amazing. Therefore 01a driving 4P1L is possible as the 4P1L doesn’t need a lot of drive. Instead of using 4P1L as a driver, we can opt for the 01a which has a similar gain. What is interesting is that the voltage swing required by 4P1L wouldn’t force the 01a outside the zone in which is highly linear, hence, with some modifications, it can work as a great driver here.

The circuit

01a into 4P1L PSE

Instead of starving the filaments of the 01a, given the voltage swing requirements for a driver, we ought to drive it at full tilt. In the circuit above, the 01a hasn’t got the stones to drive the 4P1L pair, therefore we have added a cathode follower as explained here. The M1 follower will then drive easily the output stage.

More power

Our previous west meets east circuit can be improve further. In fact, a compromise made with the filament bias design is that coupling between driver (FET follower) and the output stage wasn’t DC. We want DC coupling to get best performance, to ensure we can drive well the output stage and provide sufficient grid current even when not operating in A2.  This can be done with filament bias, however, since we are already introducing a negative supply, I’d prefer removing the filament bias and go for proper grid bias to get best performance of output stage in terms of  maximum power and linearity.

The below circuit can be easily implemented with just few modifications from previous version:

What has changed here? Not much, the coupling cap C2 is now between the gyrator and the FET follower. The gate bias resistor R6 provides high impedance to the gyrator load to ensure maximum performance of the 01a driver (minimum distortion given size of load). Not as good as previous version, but good enough. The R6 is connected to a potentiometer which sets the bias voltage. The bias voltage is derived from V2, the -50V negative supply. You can see that this circuit will put more stress into the M1 FET as now there is an additional 25V of drop across it so power burned on this device increases.

The output of the follower is directly coupled (DC) to the output stage. The filament bias resistors are removed and we use the Coleman regulators directly on the filaments of the 4P1L.

This amplifier responds better to the grid current of the output stage once the output power goes over 3.5W. At 4.5W the distortion is just above 3% (3.2%) with a 3Vpp input signal. A tad more and you can get to the 5W and a bit more into A2 operation.

I’m not a big fan of rolling valves. Perhaps it’s likely to do with the fact that I don’t have too much free time these days. However, I do look into burning in the valves. A noticeable change found on my 4P1L preamp after 350-400 hours use. The microphonic noise reduced to a minimum whilst the sound became more rounded. I added an electronic clock (LCD module) to the HT supply to monitor the exact number of hours between any changes I made to the system. It’s very handy. I’ve been running now the same pair of 4P1L valves for over 850 hours and they sound better than before. No further mechanical expansion noise is heard during warm up, something which was noticeable at the early days of use.

The second aspect I’m also in constant monitoring is the impact of the filament starvation in the 01a. Running them at 20% less current than expected is not recommended as the filaments will not operate at expected target temperature. However, after some years of joy, I’ve not seen any issues. I may remove and trace the pair of 01a in use currently to check their health.

Have fun!

I have used Jakeband teflon sockets for over 3 years. They are very well made and of high quality. Luciano from Jakeband can provide you with any custom socket you may need. This time, I requested a set of sockets for my 813 transmitting valves, a pair of UX-4 for the 300B and a pair of octal sockets for the driver stage.

Construction overview

Jakeband uses the following materials for the socket:

1. Tellurium Copper (CuTe)  for pins
2. Virgin teflon made in Germany

Jakeband sockets

The socket pins are machined with CNC from a solid core copper tellurium CuTe with tolerances within +/- 0.01 mm. The contact surface which is obtained is the highest possible.Thus there is a better grip and and a lower electrical resistance and therefore better heat dissipation (e.g. on the filament pins).

The pins undergo three surface treatments:

1. 1° electrolytic acid copper   to remove any impurities
2. 2° electrolytic silver 99.999% thickness 20 microns
3. 3°  24k gold thickness 3 microns to prevent oxidation and give more surface hardness

Why using CuTe pins?

The copper has a conductivity rating of at least 100% IACS (International Annealed Copper Standard). Brass has a conductivity rating of 28% IACS. Tellurium copper(CuTe) contact pin provides up to 320% greater conductivity than a standard brass pin

Pin gold plating

The gold plate is 3 micro-inches thick. It is purely there to prevent oxidation (and increases the surface hardness) however doesn’t  contribute to the conductive process. In fact gold is less conductive than tellurium copper. The gold plating is direct, nickel free.

Threaded pins

The new sockets come with threaded pins. This is an interesting concept that Jakeband is introducing and will be keen to test various connectors which will simplify the wiring to the socket.

I hope to test the new sockets shortly.

Contacting Jakeband

If you want to request them, you can reach out directly to Jakeband using the following form:

A minor update that is worth mentioning to everyone here. After some abuse on the J310 JFET in the gyrator board for using in a 20-30mA stage (e.g. 4P1L) I managed to kill it after a full turn-on (via switch) of the HT supply. If your supply doesn’t increase voltage somehow gently, you may want to consider this additional protection when using the J310. The solution is, as described in the Build Guide, to add any 15-18V Zener you have at hand between drain and source and using the J1 (2SK170) pads as shown below. You will have to use a low power Zener to ensure the leads fit through these pads:

Adding extra protection to the J1 FET (J310 or similar)

As part of improving my bench test gear to do sweep tests and impedance measurements, I ended up building a great preamp and buffer gig based on the SSM2019 device as described previously here.

Here is the main circuit for the preamp:

SSM2019 preamp

The circuit is same as described before. I added a rotary switch to select gain from 0dB to 60dB (ignore the diagram labels). The circuit has a DC input (differential floating) and an AC input for voltages less than 60V. There is a switch for AC/DC selection and also a switch to ground the negative input for DC mode when we don’t want it floating.

The preamp AC output is not shown but is a 1uF with a 220k resistor to ground.

The buffer circuit is similar to the above but without AC circuit and no gain selection.

Build

I used a small aluminium box to fit the whole thing. I drilled both sides to fit the BNC connectors, switched etc. A bit tight but all fit:

The circuit was built on 3 breadboard PCBs: one for the buffer, one for the preamp and one for the AC coupling circuit:

 Testing

The differential preamp jig works great. Here are some tests:

Starting with the sound card loopback as a reference, the 1kHz distortion is as low as 0.0021%:

Sound card loopback mode 0dB

When we look at the buffer output:

0dB buffer output

H2 distortion is increased by 8dB. Rest of harmonics remains the same roughly. Overall THD has risen now to 0.0033%.

If we use the AC coupling input distortion is increased slightly;

Preamp 0dB AC input

2dB for H2 and step increase for H3. Overall THD up to 0.0043% or 0.001% increase.

Now, let’s look at the buffer feeding the preamp configured at 0dB gain:

Buffer + 0dB Preamp

Overall THD is still very good and just increased to 0.0046%.

Noise floor tests

If we short the input we can look at the noise floor of the preamp in the various gain modes:

0dB preamp gain noise floor

20dB preamp gain noise floor

40dB preamp gain noise floor

60dB preamp gain noise floor

This is beautiful. Noise floor is down to less tan 640nV for 60dB gain. The workbench noise is filtering through as well as the MiniPC USB power supply ripple through the USB. With a well filtered and floating DC supply for the Sound card interface this can get even better!

Now if we couple the input in AC mode, noise floor can be as low as 140nV:

60dB preamp gain noise floor (AC coupling)

I think I have now what I needed to improve the bench measurements!

71a DHT Preamp (2012)

More than 4 years ago I ran a lovely 71a preamp which sounded amazing. I used it for some time and enjoy its sound up until I continued with my exploration around DHT preamps. Recently I was asked about how to implement this lovely valve again.

The CX371a / 71a valve is a great candidate for a line stage with its low mu and anode resistance. In my experience you have to run it above 20mA and over 100V to get the best out of this valve:

CX371a curves

The implementation of this preamp is dead simple and a few components are needed on top of the gyrator PCB:

I haven’t starved the filaments as I found this valve not to be microphonic. If you have an 01a preamp you can modify it slightly. The interesting thing is that you can run it with just 180V. Even 150V should work and you need 25mA on each channel. A J310 or BF862 lower JFET device will work fine and you will need a heatsink for the top device (e.g. DN2540). Filament resistor is anything close to 50Ω. I used some 51Ω Russian NOS wire wound resistors, but any combination will be fine.

Enjoy

Ale

An IHT preamp, oh yes!

I always loved the 27 valve. It was one of the first line stages I built many years ago before adventuring in the DHT world. I still have a large collection of them and I was very fond of the mesh anode ones. Please check Thomas’ blog in which he wrote a very nice note about it.

With the hybrid mu-follower (a.ka. gyrator) configuration, we can build a minimalistic and great preamp stage. The 27 has a mu of 9, so in some scenarios this may be a bit too much gain, but for many cases, it’s just what we need to drive the valve amps. Someone recently asked me for help on this, so here it goes my version:

The circuit is dead simple. The 27 is biased with a battery via a grid leak resistor (R1). C1 blocks DC from input and contributes to LF response by forming a pole with R1. 150nF is good enough but if you don’t have any, use 220nF. The operating point is 6mA looking at my old notebook. The supply doesn’t need any funky regulation, and 180-200V should do. The top FET should be either DN2540 or any other depletion of your choice. The lower JFET should be either a 2SK170GR or 2SK170BL (preferably). You can use a J310 here as well (or SMD BF862).

The sound is beautiful and THD is very low driven by H2 only, as you would expect from this triode.

If you don’t want battery bias, you can add a 1K5 resistor in the cathode with its decoupling cap and remove the battery and C1. R1 should be changed to 47k then.

Hope you enjoy this!

Ale

It’s been a long time since I haven’t tweaked my speakers. After more than 9 years I decided to change the drivers after falling in love with the Alpair 10M from Mark Audio. I listened to my friend Andy’s system (4P1L PSE driving the Alpairs) and decided to get hold of them.

A simple upgrade

As I don’t have much time left for DIY audio these days, I needed a simple solution. I couldn’t build a new set of speakers despite the love I have for some horn-type designs. Bringing new speakers was out of the question, so I had to modify my existing boxes to replace the FE167E. Sadly they didn’t fit straight on, so my friend Tony made me a pair of adapter boards to fit these. Made of MDF I painted them in grey:

Altair 10M Gen3 with FT96H horn tweeter

The build process took less than I expected. When I tested I noticed the treble to be higher than normal. I knew that some tweaking of the crossover was needed, in particular the level of the FT96H horn tweeter:

First test – no modification to crossover network

Indeed there was a need to attenuate the horn tweeter. I added an 8Ω wirewound L-PAD attenuator as shown below:

Crossover board (below) with attenuator (top)

Calibration was very easy to do using REWL. A few sweeps were required to get the level about right. The remaining tweaking will be done through listening:

The results are good. Very flat response between 50 Hz up to 20kHz and above. The Alpair has a higher Fo=38Hz than my previous FE167E.  I have a subwoofer so need to adjust the LF crossover to provide a better response below 50Hz:

Now, need to listen to these drivers. Allow them to settle in properly and provide some listening impressions. So far so good, the detail is impressive.

4P1L gyrator test mule

It’s always great to come back and revisit a great design. The 4P1L preamp performs flawlessly so I tweaked the gyrator board to see how it worked with the BF862 FET. The result is great, it sounds as good as it measures:

The 4P1L is biased to 150V/25mA which is the maximum current that the BF862 can do (IDSS max). You can see that the frequency response is flat up to 1.5MHz. The LF response of my test mule is affected by the AC coupling of the measuring gear. However it should be around 5-10Hz.

Introduction

Recently I was asked about whether I could write on my blog about how to design a filament bias stage. My immediate answer was around 1) I don’t have much time these days am afraid and 2) Thomas Mayer has written about it (see here). Of course, I completely forgot that Thomas never completed his intended series of posts around filament bias, so I decided to attempt explaining the practical aspects of its design in this blog.

Before you continue reading this post, I suggest you read first Thomas’ article above and get yourself acquainted with DHTs and triode amplification. I’m not going to cover any of that theory which I will give it for granted that the reader is experienced with valve circuits and in particular with the hybrid mu-follower amplification stage with gyrator load.

3A5 DHT example

Long time ago I played around with the 3A5 DHT. This lovely sounding valve has a high mu for a DHT and also its filaments aren’t current demanding so it’s a nice candidate for filament bias.

When I tested this valve I found that it performed really well at low level signals and with both triodes connected in parallel. At Ia=10mA / Vak=100V distortion was minimised and sounded really nice. We will start with that point as our target operating point for this example.

This valve has a highish anode resistance around 8.3KΩ and 1.8mA/V transconductance with a gain (mu) of 15 when you look at the specifications. When I traced it, my sample had a lower mu around 12.4-13. When triodes are in parallel the transconductance will double, voltage gain is maintained so anode resistance also is halved. My DUT with both sections in parallel delivered about less than 5K of anode resistance and 2.5mA/V of transconductance at the selected operating point. Mu was about 12.45. With a gyrator load, this triode is a nice candidate for a preamp stage given the x12 – x15 (or 22-23.5dB) gain that can be achieved as well as decent anode current which will avoid any slew rate issues when driving the next stage (the main amplifier for example). The mu output of gyrator will deliver low impedance which is great.

When we look at the triode curves of both sections in parallel, we can see that the grid to cathode voltage (Vgk) required for the Ia=10mA at a anode to cathode voltage (VaK) of 100V is somewhere around -2 and -3V curves. In fact the point is -2.3V:

Note: These curves are generated by the Spice model derived from traced curves

The actual load of the triode with the gyrator is the output load in parallel with the bootstrapped mu resistor from the gyrator which presents a very high impedance. Therefore, the actual reflected load is simply the 100K input impedance of next stage in this example.

In filament bias, we want to elevate the cathode DC level to the bias point we want. For that we have to take into account the following points:

1. Filament current (IF)
2. Cathode current (IK)
3. Target voltage (VGK)

The filament resistor needed (RFIL) is VGK/(IF+IK). This is due to the fact that the anode current which is also the cathode current in a triode, adds to the voltage drop across the resistor. In practice, the valve parameters vary greatly (+20%) so we don’t need to be very accurate here. In fact, if we use a gyrator load, the anode voltage and anode current can be adjusted by the gyrator CCS reference voltage.

So in our example we can derive the filament resistor by using the formula above  RFIL = 2.3V/(200mA+10mA)=10.95Ω. You can be precise and look for a 11Ω resistor by combining 10Ω and 1Ω ones. I will stick to 10Ω simply because we can use one resistor and the variance will be minimal. One point to consider is the power dissipated in the RFIL. This is the killer in the filament bias circuits. In this case, thanks to the low power filaments and currents in place, the Pd (RFIL) = (IF+IK)^2 * R. We want at least x2 Pd capability on the resistor. I have a 5W wire wound at hand, so will use that one.

Sometimes you will find that there can be an offset voltage in your calculations/simulations due to the way the triode curves have been plotted. The ones from the data sheet are typically for AC heater or DC heaters with cathode referenced to mid point via a pot. I generally trace my curves with the negative filament to ground. So the difference you may experience in anode voltage may be due to the way the elements are connected and how the original curves were derived:

3A5 DHT preamp using gyrator load

So If we set the gyrator voltage to about 105V like above, the actual Vgk will be 103V if we subtract the bias voltage which is the reference for the triode.

The above circuit performs really well with a very low distortion for low level signal (THD<0.1% @1Vrms). I measured less than 0.3% @10vrms, so we should expect really low distortion. The current drive is excellent at 10mA so it can take some heavy loads. The frequency response is excellent and flat above 1Mhz so we should expect some oscillation potentially if we don’t add the proper stopper resistors and or ferrite beads.

Just bear in mind that if you build yourself the 01A Preamp Gen2, you can adapt it very easily to fit the 3A5 with minimum modifications.

Hope you found this example clear enough for your to embark in designing the filament bias yourself. A simple and delightful experience in my view!

It was pure lust and love at first sight. I found the RL12P35 german transmitting pentodes and couldn’t resist in buying them. A very nice set of NOS Telefunken and Valvo valves with also NOS sockets. These pentodes look like a de-rated LS-50/GU-50 and very interesting candidates for a nice PP amp:

These have been used by Lorenz to build the classic and lovely PP amp LVA/BA30 RL12P35 Amp with anode to grid feedback (i.e. a la Schade) like the below amplifier:

When I submitted to the mercy of the uTracer, I had to work around the 300V limitation of my uTracer v3 (I do have the 400V upgrade kit but I haven’t got around to modify it yet). So I looked into the data sheet to find an operating point to measure my valves against:

RL12P35 test point

The valves measured really well, most of them above 90%. The test point used was Va=250V, Vs=200V, Vgk=-15V which aimed for Ia=110mA:

RL12P35 (valve 9) used for creating the Spice Model

From my traces I could see that the screen current should be about 25mA, Ra closer to 24-25kΩ and transconductance around 4.5mA/V which is predicted from the data sheets.

The generated Spice model matches really well:

RL12P35 @Vs=250V

RL12P35 @Vs=200V

RL12P35 Screen Current @Vs=200V

RL12P35 triode curves

Here is the Spice model of the pentode: RL12P35-pentode-150V:

****************************************************
.SUBCKT RL12P35-pentode 1 2 3 4 ; A G2 G1 C;
* Extract V3.000
* Model created: 28-May-2016
*
* Traced and model created by Ale Moglia / valves@bartola.co.uk
* www.bartola.co.uk/valves
*
*
*
X1 1 2 3 4 BTetrodeDE MU= 5.01 EX=1.331 kG1= 676.1 KP= 14.9 kVB = 1008.8 kG2= 3519.4
+Sc=.12E+00 ap= .034 w= 16. nu= .24 lam= 11.8
+ Ookg1mOokG2=.119E-02 Aokg1=.46E-06 alkg1palskg2=.119E-02 be= .151 als= 3.10 RGI=2000
+ CCG1=16.5P CCG2 = 0.0p CPG1 = 0.05p CG1G2 = 5.7p CCP=10.4P ;
.ENDS

****************************************************
.SUBCKT BTetrodeDE 1 2 3 4; A G2 G1 C
*
* NOTE: LOG(x) is base e LOG or natural logarithm.
* For some Spice versions, e.g. MicroCap, this has to be changed to LN(x).
*
RE1 7 0 1MEG ; DUMMY SO NODE 7 HAS 2 CONNECTIONS
E1 7 0 VALUE=
+{V(2,4)/KP*LOG(1+EXP(KP*(1/MU+V(3,4)/SQRT(KVB+V(2,4)*V(2,4)))))}
E2 8 0 VALUE = {Ookg1mOokG2 + Aokg1*V(1,4) - alkg1palskg2*Exp(-be*V(1,4)*SQRT(be*V(1,4)))}
E3 9 0 VALUE = {Sc/kG2*V(1,4)*(1+tanh(-ap*(V(1,4)-V(2,4)/lam+w+nu*V(3,4))))}
G1 1 4 VALUE = {0.5*(PWR(V(7),EX)+PWRS(V(7),EX))*(V(8)-V(9))}
G2 2 4 VALUE = {0.5*(PWR(V(7),EX)+PWRS(V(7),EX))/KG2 *(1+als*Exp(-be*V(1,4) * SQRT(be*V(1,4))))}
RCP 1 4 1G ; FOR CONVERGENCE A - C
C1 3 4 {CCG1} ; CATHODE-GRID 1 C - G1
C4 2 4 {CCG2} ; CATHODE-GRID 2 C - G2
C5 2 3 {CG1G2} ; GRID 1 -GRID 2 G1 - G2
C2 1 3 {CPG1} ; GRID 1-PLATE G1 - A
C3 1 4 {CCP} ; CATHODE-PLATE A - C
R1 3 5 {RGI} ; FOR GRID CURRENT G1 - 5
D3 5 4 DX ; FOR GRID CURRENT 5 - C
.MODEL DX D(IS=1N RS=1 CJO=10PF TT=1N)
.ENDS BTetrodeDE

Enjoy!

Ale

A very interesting Russian directly-heated pentode related to 4P1L is the 2P29L. It has a similar mu (μ=9), much higher anode resistance 2.8-3KΩ and transconductance of 3mA/V when triode-strapped. The filament requirements are much lower at 120mA. I picked one valve from my collection to submit it to the mercy of the curve tracer:

2P29L test point (pentode)

The triode curves are really nice:

2P29L triode curves and model

This valve is as linear as the 4P1L (hooray). As a preamp it can be easily implemented like the 4P1L Gen2 preamp using a gyrator PCB which simplifies the building process:

2P29L preamp

Running it at 15mA and slightly above the recommended 160V achieve its lowest distortion.

We could also use this valve as a driver for a 4P1L preamp, which comes very handy for filament bias:

Looking for more

Well, ok. You want more power from previous 4P1L PSE amp? Here is an alternative approach:

2P29L into 3x4P1L PSE

You can get 5W at 2% THD maximum. I’d rather avoid filament bias at the output stage and instead apply fixed bias and a source follower driving the 4P1L stage. Best performance, lower THD at high output power. However, the above circuit is dead simple to implement without adding extra complexities!

Introduction

More than 5 years ago, I built a fantastic single-ended amp with the unique 45. The 45 has a distinguished tone and personality despite its mere 2W of output power. If you have high efficiency speakers, then it’s a great amplifier to build. With 2W you can enjoy music in a mid-sized room. You don’t need more, this amplifier performs at its best at low output levels and in particular when playing jazz or classical music.

The 45 Amp design

There are plenty of design circuits out there. I settled for a simple triode driver using a gyrator load. The choice was down to the 6J5 and 7193 (a military version of the 2C22). Both triodes are medium mu and sound really nice. Depending your needs, you may opt for a different driver (even pentode). However, they need to be able to drive the large voltage swing required by the 45. I’d go for a 6J52P, 6e5P, 6e6P, D3a or C3g these days. It all depends on your needs and available valves. The driver is biased at 7mA to provide enough grid current to avoid slew rate issues. An improved version would be to add a MOSFET follower to provide better performance under grid current. An example of a follower implementation can be found here.

The 45 is biased hot at 34mA/300V. The anode can handle 10W and this operating point provided best sound in my view. The OT is crucial, so invest as much money as you can afford. Rod Coleman regulators are needed to implement this amp without hum and the unwanted  inter-modulation effects.

I carried out several tests on the driver to find the sweet spot for minimum distortion and full swing. The driver is a hybrid mu-follower composed by the gyrator and the 6J5/7193 triode. The valve is biased by a set of 5 red LEDs to about 8V. I think I had a combination of a white LED and LED to provide 8V in my implementation. The dynamic resistance is minimum and won’t impact the performance of the stage.

I used the Sylvania metal-base 6J5 but then settled for the 7193 valves. They sounded better and I was quite pleased with the overall performance of the amplifier.

The amplifier design is very simple. With the gyrator PCB you can simplify the driver build and also use different valves to experiment with them. I originally didn’t have a PCB so I built my gyrator in a prototype board.

45SE Amplifier upgraded with the 7193 drivers

I’d highly recommend you building this amplifier. If you want to experience the single-ended sound, then this is one of the amps to build. Of course you can go for higher power levels with a 2A3 or 300B, however, the sound of the 45 is unique. Worth trying

IN1166 DUT

This is a long overdue post, as I have carried out these tests time ago but never got around to publish the results. Dorin from DvB transformers sent me a pair of his transformers for testing. These are great devices for amp input stages, particularly for a nice Spud amp using medium mu output stages with 6P15P or 6e5P for example.  Either PP or SE output, this will give you a great solution for a minimalistic design:

The configuration options are multiple, you can see what you can do with a similar transformer spec from DvB:

I did some tests without any Zobel compensation and here was the initial result:

First test without Zobel compensation

The nasty peak at 40kHz can be easily corrected by adding a 8k1 resistor in series with a 1,600pF capacitor as Zobel arrangement across the secondary:

With Zobel compensation

The transformer can do a nice flat response from DC to 40kHz (-3dB) in a 1:6 arrangement as shown above. This is really good.

The distortion response is really good as well. Let’s have a look at the response of the IN1166 transformer with different input signals. Starting with a 1Vrms output:

The third harmonic is higher than the second which is due to the distortion introduced by the transformer.  However, the overall THD is very low (0.0028%). It’s rather more interesting to see its performance when swinging larger volts. Here is the example at 0.008% for 5Vrms output level

The response at 10Vrms output level is shown below. The distortion harmonic profile changed with a higher H2 and decaying H3 and H4 which is nice to see. Overall THD is 0.006% @10Vrms which is very nice to see:

Overall, a great and flexible transformer to use in my opinion. A nice cathode follower to drive it to ensure a good HF response due to leakage capacitances in play and the reflection of the Miller capacitance of the output stage into the primary which is the challenge of using a step up transformer in a Spud amp.

Contact Dorin directly if you want one of this. He can be reached on his email account which is dorin.bodea at gmail.com.

Ale